Wireless transmitters can benefit from measurement and control of RF power. Because of such factors as regulatory requirements and the need to co-exist with other wireless networks, the RF power levels of wireless transmitter high-power amplifiers (HPAs) must be monitored and controlled. The precision and accuracy of these measurements can result in improved transmitter spectral performance as well as significant savings in operating cost for an HPA.

Some form of factory calibration of a PA’s output power is usually performed as part of any scheme to regulate the PA’s output power. Calibration algorithms vary vastly in terms of their complexity and effectiveness. This article will focus on how a typical RF power control scheme is implemented and will compare the effectiveness and efficiency of various factory calibration algorithms.

Figure 1 shows a block diagram of a typical wireless transmitter with RF power measurement and control functionality. A small portion of the signal from the HPA is coupled and fed to an RF detector for measurement. The coupler is located near the antenna, and after the duplexer and isolator; their associated losses must be included as part of the calibration.

Depending upon the coupling factor, the signal from the directional coupler will be proportionately lower (such as 20 or 30 dB lower) than the signal going to the antenna. Coupling power in this manner results in some power loss in the transmit path, usually a few tenths of a dB depending upon the quality of the directional coupler. In wireless infrastructure applications where maximum transmitted power typically ranges from +30 to +50 dBm (1 to 100 W), the signal coming from the directional coupler will still be too large for the RF detector. As a result, additional attenuation is generally required between the coupler and the RF detector.

Modern logarithmic-responding RF detectors (logamps) have a powerdetection range between 30 and 100 dB, and provide a temperature- and frequency-stable output. In most applications, the detector output is applied to an analog-to-digital converter (ADC) to be digitized. Using calibration coefficients stored in nonvolatile memory (EEPROM), the code from the ADC is converted into a transmitted power reading. This power reading is compared to a set-point power level. If any discrepancy between setpoint and the measured power is found, a power adjustment, which can be made at any one of a number of points in the signal chain, will take place. The amplitude of the baseband data driving the radio can be adjusted, a variable-gain amplifier (VGA) at RF or intermediate frequency (IF) can be adjusted, or the gain of the HPA can be changed. In this way, the gaincontrol loop regulates itself and maintains transmitted power within desired limits. It is important to note that the gain-control transfer functions of voltage-variable attenuators (VVAs) and HPAs are often quite nonlinear. As a result, the actual gain change resulting from a given gain adjustment is uncertain. This reinforces the need for a control loop that provides feedback on changes made and further guidance for subsequent iterations.

In the system of Fig. 1, almost none of the components provide very good absolute gain accuracy specifications. The impact of this can be shown by targeting a transmit power error of ±1 dB. The absolute gain of devices such as HPAs, VVAs, RF gain blocks, and other components in the signal chain will generally vary from device to device to such an extent that the resulting output power uncertainty will be significantly greater than ±1 dB. In addition, signal chain gain will vary further as the temperature and frequency change. As a result, it is necessary to continually measure the power being transmitted.

Output power calibration could be defined as the transfer of the precision of an external reference into the system being calibrated. A calibration procedure generally involves disconnecting the antenna and connecting an external measurement reference, such as an RF power meter, in order to transfer or clone the meter’s accuracy to the transmitter’s integrated power detector. The calibration procedure involves setting one or more power levels, taking the reading from the power meter and the voltage from the RF detector, and storing all of this information in nonvolatile EEPROM. Using this stored information, the transmitter can precisely regulate its own power without the power meter connected. As parameters such as amplifier gain vs. temperature, transmit frequency, and desired output power level change, the (calibrated) onboard RF detector will act like a built-in power meter with an absolute accuracy that will ensure that the transmitter is always emitting the desired power within a defined tolerance.

The transfer-function linearity and stability over temperature and frequency of the system’s RF detector strongly influences the complexity of the calibration routine and the 7achievable post-calibration accuracy. Figure 2 shows the transfer function of an RF logamp with behavior versus temperature exaggerated for illustrative purposes. Three curves are shown: output voltage versus input power at +25°C, +85°C, and –40°C. At +25°C, the output voltage of the detector ranges from around 1.8 V for an input power of –60 dBm to 0.4 V for an input power of 0 dBm. The transfer function closely follows a straight line that has been laid over the trace. The transfer function deviates from this straight line at the extremities, and there are also instances of nonlinear behavior at power levels between –10 and –5 dBm.

A quick calculation suggests that this detector has a slope of approximately –25 mV/dB: a 1-dB change in input power will result in a 25-mV change in output voltage. This slope is constant over the linear portion of the dynamic range. So, notwithstanding the slight nonlinearity that was identified at around –10 dBm, it can be concluded that the behavior of the transfer function at +25°C can be modeled using a simple equation in the form of Eq. 1:

VOUT = SLOPE × (PIN – INTERCEPT) (1)

where

INTERCEPT = the point at which the extrapolated straight-line fit crosses the x-axis of the plot.

From a calibration perspective, the simplicity of this equation is useful as it will allow the transfer function of the detector to be established by applying and measuring as few as two different power levels during the calibration procedure.

Consider the behavior of this imaginary detector in Fig. 2 over temperature. At an input power of –10 dBm, the output voltage changes by approximately 100 mV as the temperature shifts from about room temperature to either –40°C or +85°C. Based on the earlier calculation of the detector’s slope as being –25 mV/ dB, this equates to a deviation in measured power of ±4 dB, which is too much deviation for most practical systems (real-world RF detectors typically have temperature drift between 0 and +/-0.5 dB). In practice, what is needed is a detector with a transfer function having minimal drift versus temperature. This will ensure that a calibration procedure performed at ambient temperature will also be valid over a wide range of operating temperatures. This allows the transmitter to be factory calibrated at ambient temperature and avoiding expensive and time-consuming calibration cycles at hot and cold temperatures.

If the transmitter is frequency-agile and needs to transmit at multiple frequencies within a defined frequency band, the behavior of the detection as a function of frequency is also important. Ideally, the RF detector should exhibit a response that does not change significantly within a defined frequency band. This makes it possible to calibrate the transmitter at a single frequency and be comfortable that there will be little or no loss of accuracy as the frequency changes.

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