Microstrip and suspended-stripline transmission techniques can be combined to create a compact electrically tunable preselector filter for L-band applications.
Electrically tunable preselectors are key elements in communications, avionics, and radar receivers. Narrowband RF and microwave preselectors prevent large off-channel signals from overdriving a receiver front end. Microstrip combline and interdigital tunable filters have been described by several authors.1-5 By combining a suspended-stripline bandpass filter (BPF), microstrip low-noise amplifier (LNA), and input/output matching networks, an electrically tunable L-band preselector can be assembled with typically 3-dB bandwidth from 18 to 24 MHz.
The tunable BPF is split to provide partial selectivity with minimum insertion loss prior to amplification for improved input noise figure. The first two-pole filter before LNA prevents undesirable signals from overdriving the LNA. The second three-pole filter after the LNA provides selectivity against receiver image and spurious frequencies. The three-pole BPF placed after the LNA has a negligible effect on the overall preselector input noise figure.
A conventional combline filter (Fig.1) consists of a set of parallel-grounded resonators loaded by the variable capacitors made of tuning screws. Combline filter can be realized on different transmission lines. Suspended stripline provides high quality factor (Q) of approximately 500, stability over a wide temperature range, and high impedance range.6 In the high-Q suspended stripline (Fig.1b), the parallel strips are printed on both sides of a substrate in a symmetrical configuration. Plated through holes (vias) provide electrical connection between top and bottom conductors. When dual-center conductors are located symmetrically with respect to each other, they are excited in phase, causing most of the electromagnetic field to propagate in the air dielectric. Therefore, substrate dielectric losses and dielectric constant variations have negligible effects on the attenuation and phase velocity of the transmission media.
Suspended stripline resonators are placed between two parallel ground planes. Adjacent suspended stripline resonators are coupled by the fringing fields. The typical length of the combline filter resonators is between Λ0/16 and Λ0/6, where Λ0 is the center guide wavelength at the resonator. For this resonator length, magnetic coupling predominates.7 The minimum practical length of the resonators is limited by the Q. Practical Q values are dependent upon the ground-plane spacing (base), the frequency of operation, the finish of the ground surface, the plating material of the printed-circuit board (PCB), and the suspended-stripline structure.
Short resonators yield a compact structure with excellent stopband performance. With a resonator length, l, of l =Λ0/8, the second passband will appear at better than four times the fundamental operating frequency, while at l = Λ0/16, the second passband will occur at more than eight times the fundamental frequency. Combline filter trade-offs for various resonator lengths are described in ref. 4.
The bandwidth of a combline filter is a function of the ground-plane spacing, b, to wavelength ratio, b/Λ0 and the spacing, S, between resonators. The bandwidth increases with higher S and b/Λ0. For combline filters, bandwidths of 2 to 50 percent can be achieved.
The spacing (b) between two ground planes (cover and housing) defines resonator impedances and lengths and maximum power and Q. Resonator impedances range from 70 to 140 Ω at frequencies (f) of f < 1 GHz. Large bases (ground planes) lead to higher power handling and increased Q, but also to an increase in resonator length and housing height.
Spacing S (between resonators) is proportional to base b. For a distance between printed resonators and ground planes of b/2 = 200 mils, with a substrate thickness (h) of 10 mils, the base should be equal to b + h = 410 mils. For these conditions, the resonator impedance is approximately 100 Ω (an admittance of 0.01 Ω−1.
The loading capacitance for each combline resonator is1:
where:
YI = the admittance of the ith resonator when the (i − 1)th and (i + 1)th resonators are shorted and θ0 = (2πl)/θ0 = the electrical length at the center frequency.
For 1-GHz suspended-stripline resonators with the above dimensions, the guide wavelength is equal to θ0 = 28.8 cm. According to Eq. 1, the total capacitance, CTOT = 2.75 pF for a resonator length of θ0 = 30 deg. (l = Λ/12).
Usually, capacitors are also used as tunable elements to compensate for production tolerances. The use of capacitors is especially critical for narrow bandwidth. At low frequencies, capacitor Q's are higher than the Q's of resonators. At microwave frequencies and for higher capacitance values, capacitor Q's can be lower than resonator Q's, dominating performance when filter losses are calculated.
Table 1 compares experimental results for tunable combline filters with Λ/12-long suspended-stripline resonators and with air-dielectric trimmer capacitors (Giga-Trim products from Johanson Corp., Boonton, NJ) with capacitance range of 0.4 to 2.5 pF.
Figure 2a shows an electrically tunable BPF consisting of suspended-stripline resonators grounded at one end, high-Q GaAs varactor diodes, and lumped-element loading capacitors between the ground plane and the other end of each resonator. The tunable BPF was realized with reverse-biased varactor diodes D1, D2, D3, D4, and D5 which were used as tuning elements to adjust the combline passband over the full frequency range. Tuning is performed by altering the bias of the varactor diodes. A two-pole BPF fabricated with trimmers provided a 3-dB BW of 41 MHz with 1.2 dB insertion loss. A similar two-pole BPF filter with varactors yielded a 63.8-MHz BW with 1.85-dB insertion loss.
Preselector selectivity depends on filter Q. The total Q of the tunable BPF is taken as the combination of the resonator, loading capacitor, and varactor diode Q's. For a low-loss L-band combline filter, the varactor diode Q is an very important parameter. The Q of the best varactor diodes is lower than that of the suspended stripline resonators and loading capacitors, and is a dominating factor when calculating filter losses.
To increase diode Q, GaAs varactors can be used. For example, model MA46617 GaAs abrupt varactor diodes provide a Q of 210 at 1 GHz and serve as effective variable capacitors. For the MA46617 diodes, the total capacitance ratio, CT0/CT45 = 4.4 to 6.9, where CT0 and CT45 are the varactor capacitances at 0 and 45 V, respectively. Varactor tuning voltages are controlled by a microprocessor to precisely tune the filter response.
The total loaded capacitance of combline tunable filter is given by:
where:
Cj = the varactor diode junction capacitance and
C = the parallel lumped-element capacitance (C1 = C2 = C3 = C4 = C5 = C). The center frequency of the tunable preselector is determined by lengths of resonators and the total loaded capacitance, CTOT.
For the 1-GHz tunable filter with resonator length of l = L/12, the total capacitance (from Eq. 1) is CTOT = 2.75 pF. For a 25-percent tuning range, varactor diodes with a junction capacitance of Cj = 1.3 pF at −4 VDC can be used. In this case, the lumped-element capacitance (according to Eq. 2) should be C = 2.75 − 1.3 = 1.45 pF. To allow for biasing, capacitors C6, C7, C8, C9, and C10 are added (Fig. 2a). These capacitors provide an RF short for the varactors and an open circuit for the bias currents.
The tunable preselector is based on a combination of suspended-stripline and microstrip transmission lines. Each of these two transmission-line formats offers certain strengths.6 Microstrip, for example, supports a high level of integration and excellent heat dissipation, since a good ground connection (with minimal reactance) is needed for each device. The tunable preselector includes a microstrip LNA between the first and second suspended-stripline BPF. The LNA is based on a monolithic-microwave-integrated-circuit (MMIC) MGA-85563 device from Agilent Technologies.8 The RF layout of the LNA is shown in Fig. 2b.
The LNA operates from a +3-VDC bias supply and draws nominal current of 15 mA. For the lowest noise-figure performance, the amplifier's input port should be matched with the output of the two-pole combline filter and the input of the three-pole combline filter. To match the input of the LNA to the 50-Ω two-pole BPF output, inductor L1 is placed in series with the input of the LNA. DC blocking capacitor C12 is placed at the output of the MMIC LNA to isolate the amplifier from the three-pole BPF. Inductor L2 and capacitor C13 isolate RF from the DC supply.
A circuit that includes the MMIC LNA and its input and output matching networks was realized on a microstrip line using the same BPF dielectric substrate. The combine-filter input/output network provides a transition from the microstrip-line LNA to the suspended-stripline BPF as well as matching between the LNA input/output ports and the suspended stripline resonators. For low-frequency application (less than 1 GHz), a straight connection between microstrip and suspended stripline can be used. This connection includes the step between center conductors of two lines and ground plane step (Fig. 2c) to provide a 50-Ω impedance for both lines. For higher-frequency applications, the special transition between the two lines can be used.6
Matching sections at the filter input and output match the resonators with the 50-Ω microstrip lines. Each matching network (Fig. 3) consists of two high-impedance suspended-stripline series printed inductors (L1 and L2) and low-impedance suspended stripline (shunt capacitance Cm). This matching network is equivalent to the T-section of the LPF (Fig. 3b).
The main parameters of the matching network can be determined using the wave-matrix method. This matching circuit is a two-port network, which is equivalently represented in the form of four cascade-connected elementary two-port networks (Fig. 3b). The resulting transfer matrix of the equivalent two-port network is equal to the product of the transfer matrices of the above component two-port networks written down in the order of energy flow.
Multiplying the transfer matrices for the center frequency yields:
For the case of perfect matching of input port 1, element S11 of the scattering matrix, which has the physical meaning of reflection coefficient, must be equal to zero, S11 = T21/T11 = 0; therefore, T21 = 0. If Z1 = Z3 = Z, it is possible to obtain from Eq. 3:
From Eq. 4, it is possible to find the value of the capacitance matching element and its dimensions.
Using these techniques, the preselector was fabricated on 10-mil-thick dielectric substrate TLE-95™ from Taconic Advanced Dielectric (Germantown, NY) with dielectric constant of 2.95. The PCB was suspended over a silver-plated aluminum machined housing. The depth of the housing and cover was 0.200 in. (0.508 cm). The total dimensions of the preselector are 12.7 × 5.08 × 1.27 cm.
Figure 4 shows the frequency response of the preselector for various varactor tuning voltages. As the filter tunes from 962 to 1213 MHz, its 3-dB bandwidth varies from 18.1 to 24.4 MHz. Table 2 summarizes the preselector's performance.
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