Download this article in .PDF format
This file type includes high resolution graphics and schematics.

System performance is often limited by the available antennas, whether the system is a radar or communications network. A reflectarray antenna represents an alternative to conventional reflector antennas and can provide high gain, but is typically limited in bandwidth.1 It consists of a feed (usually a pyramidal horn antenna) and a flat multiple-element reflector formed of reflectarray elements. These elements form a plane wave in a desired direction by modifying the phase of signals impinging upon them from the feed. Reflectarray antennas offer several benefits; these include high gain and the potential for beam-scanning applications, as well as the facts that it can be fabricated with standard photolithography and can be designed without a heavy beam-forming network. Of course, a reflectarray antenna has one major disadvantage that must be overcome to apply it in practical systems—narrow bandwidth.2


Related Articles
Helical Antenna Links GSM/UMTS
Phased-Array Antennas Aid Wireless Communications
Compact Automotive Antennas Juggle Multiple Bands


A reflectarray antenna’s bandwidth is limited by two factors: differential spatial phase delays3,4 and the limited bandwidths of single microstrip reflectarray elements. Phase delays can be minimized by choice of reflector size and distance between the feed and reflector and by optimizing the phases of signals retransmitted by certain parts of the reflector.4 Differential spatial phase delays can be minimized to the degree that the reflectarray element becomes the main limiting factor in determining the bandwidth of a reflectarray antenna.

To explore the wider bandwidth potential of a reflectarray antenna, a microstrip reflectarray element was developed with relatively wide bandwidth (18% for VSWRs of less than 2.0:1) and high front-to-back ratio. The element also features highly linear phase tunability versus transmission line length (to help minimize differential spatial phase delays among antenna elements) and minimizes the sensitivity of an antenna to manufacturing tolerance errors.

 

1. These sketches show the signal directions for a (a) standard transmit-receive antenna and (b) reflectarray antenna element.

To better understand the characteristics of the reflectarray elements, they were first designed as part of a regular transmit-receive antenna [Fig. 1(a)].2 This design includes an input port connected to one end of a transmission line and is very much a conventional microstrip antenna. A number of methods are available for increasing the bandwidth of a microstrip antenna, such as making different cuts and slits in the patch,10,13 loading the patch with chip elements,9,11 using multiple stacked patches,7,8 or using parasitic multiresonantor configurations.12,14 One of the most effective ways of increasing the bandwidth of a microstrip antenna is by using the aperture coupled method of feeding energy to the patch.5,14 Microstrip antennas that employ this type of feed typical feature a bandwidth in excess of 8%5 while also achieving a low geometrical profile, although such broad bandwidth is usually not available with a conventional microstrip antenna. Figure 2 offers a schematic diagram for an aperture coupled antenna.

2. These diagrams show (a) a conventional aperture-coupled antenna and (b) the modified aperture-coupled antenna.

In addition to wideband operation, an aperture-coupled antenna provides good isolation between the patch and feed network, which translates into low spurious radiation of the feed lines. This also allows a designer to choose different substrates for the feed network and the patch, not requiring the patch and feed network to be placed on one plane. With these benefits, aperture-coupled antennas are frequently used as radiating elements for phased-array antennas.17

With some modifications, it is possible to improve the performance of aperture-coupled antennas. These modifications can yield wider bandwidths, higher front-to-back ratios, and tigher cross sections. To evaluate the effects of the modifications, two different antennas were designed: one with and one without modifications [shown schematically in Figs. 2(a) and 2(b)], both working at S-band frequencies (f0 = 3.4 GHz). The first antenna is a regular inverted-patch, aperture-coupled antenna (IPACA).5,14 It was constructed with a thick (h = 0.07λ) layer of ROHACELL® HF foam from Evonik Industries placed between the patch and slot.15 According to work reported in refs. 9 and 16, a thick layer of dielectric material reduces the quality factor (Q) of a patch antenna, increasing its bandwidth. In addition, with the low dielectric constant of the foam (1.06), surface waves are less likely to occur. The coupling slot has dimensions of a = 27 mm and b = 2.7 mm, with the 1:10 dimensional relationship chosen per ref. 5. To reduce the back radiation and to increase efficiency, a ground plane was used at a quarter-wavelength distance from the feed line. The feed line length was modified to compensate for the slot reactance.5 In practice, the quarter wavelength of the feed line suggested in ref. 5 turns out to be much smaller. For this first experimental antenna, this length amounts to bstrip = 10 mm, or 0.15λ in the transmission line.

3. These performance plots show (a) the VSWR for a standard aperture-coupled antenna as well as its radiation patterns in the (b) E-field cut plane and the (c) H-field cut plane, with cross-polarization shown in blue.

The aperture-coupled antenna was analyzed with the aid of CST Microwave Studio 2011 electromagnetic (EM) simulation software from Computer Simulation Technology. Figure 3 shows the results of the analysis, revealing a bandwidth of 7% for VSWRs of less than 1.50:1 (from 3.32 to 3.57 GHz) or a bandwidth of 12% for VSWRs of less than 2.0:1. It should be noted that for an X-band inset-fed microstrip patch antenna, bandwidths reach 5% for VSWRs of less than 2.0:1.6

The radiation pattern of a standard aperture-coupled antenna is similar to the radiation pattern of an inset-fed patch antenna. The gain of the aperture-coupled antenna amounts to 8.9 dB at 3.4 GHz and does not drop below 8.8 dB throughout the bandwidth of the antenna. The cross-polarization level in the E-field plane is less than -30 dB for angles from -30 to +30 deg. (and less than -28 dB for the full cut plane). For the H-field cut plane, the cross-polarization level is less than -28 dB for the full cut plane. The highest cross-polarization level can be found in the φ = 45 deg. and φ = 135 deg. cut planes at -16 dB. The front-to-back ratio (defined as the ratio between the power radiated into the front hemisphere of the antenna and the power radiated in the opposite direction) is a maximum of 25 dB, and does not drop below 24 dB within the antenna’s 7% bandwidth. The total antenna efficiency reaches a maximum of 96%, staying above 93% for the full 7% bandwidth of the antenna.

The author’s modifications[see Figs. 2(b) and 4] resulted in improvements in bandwidth, efficiency, and front-to-back ratio. The most important modification was using two identical, parallel coupling slots. These slots have a length of dslot = 23 mm and width of bslot = 1.6 mm and are cut symmetrically about the center of the patch. Compared to the structure of the standard aperture-coupled antenna [Fig. 2(a)], these modified slots are almost two times thinner, so that the coupling aperture is divided. This in turn has a positive influence on the front-to-back ratio. In the modified antenna design, the front-to-back ratio reached a maximum value of 27 dB, staying about 23 dB across the 18% bandwidth of the antenna. Another modification was the use of an U-shaped transmission line to enable the slots to be fed more uniformly.22 According to refs. 8 and 23, uniform slot excitation means higher coupling between the transmission line and the patch, leading to  higher total antenna efficiency. For the modified aperture-coupled antenna, the total efficiency reaches a maximum value of 98%, staying above 93% throughout the antenna’s 18% bandwidth.

4. These performance plots show (a) the VSWR for the modified aperture-coupled antenna as well as its radiation patterns in the (b) E-field cut plane and the (c) H-field cut plane, with cross-polarization shown in blue.

These modifications of an aperture-coupled antenna, the use of a double parallel slot and the U-shaped transmission line, resulted in a frequency range of 3.17 to 3.65 GHz, or a 14% bandwidth for VSWRs of less than 1.50:1. For VSWRs of less than 2.0:1, the bandwidth amounts to almost 18%. Figure 4 shows these parameters as analyzed via a CST Microwave Studio 2011 simulation.

Download this article in .PDF format
This file type includes high resolution graphics and schematics.

Figures 4(a) and (b) show that the radiation pattern of a modified aperture coupled antenna is similar to the radiation pattern of a standard inset fed patch antenna. The antenna gain for the modified structure amounts to 8.9 dB at 3.4 GHz, remaining above 8.7 dB throughout the 18% bandwidth. The E-field cross-polarization level is less than -30 dB between -30 deg. and +30 deg. (and less than -28 dB for the full plane), while the H-field cross-polarization level is less than -28 dB for the full cut plane, with levels as high as -16 dB at φ = 45 deg. and φ = 135 deg.

The phase shifts brought about by the microstrip reflectarray elements [Figs. 2(a) and (b)] are adjusted by changing the lengths of transmission lines. To find required transmission-line lengths, the input port [Fig. 1(b)] from the designed microstrip antenna [Fig. 1(a)] is removed.2 As part of the design process, the phase shift and attenuation of the retransmitted signal (from the feed) are examined. It is usually assumed that a reflectarray element is surrounded by an infinite number of identical elements (infinite periodic array).2,24 In this way, it is possible to approximate the behavior of a reflector formed of a finite number of elements.

5. This sketch depicts a reflectarray antenna element inside a rectangular waveguide.

A popular analysis method based on this assumption is the Waveguide Simulator Method,4,18,19 in which part of a reflectarray reflector is placed inside a rectangular waveguide for analysis (Fig. 5). Inside the waveguide, the reflector is illuminated by a wave propagating along the waveguide. In this way, it is possible to simulate reflector illumination by a wave impinging from a direction calculated by Eq. 118,20,21:

where:

a = the width of the waveguide.

For TE10 transverse-electromagnetic propagation, mode angle θ refers to an angle that is defined in the H-field plane. It means that a single waveguide simulator can simulate reflector illumination by a plane wave from a certain direction and for a certain frequency. For the reflectarray elements based on antennas designed in point 2 a Waveguide Simulator analysis was carried out, using CST Microwave Studio 2011 (Fig. 6).

Figures 5(c) and 6(a) show the phase shifts of retransmitted signals versus length of transmission line, for the reflectarray elements from both antenna designs. For the modified design, analysis was performed for a wider bandwidth. Comparison of Figs. 6(a) and (c) reveals that the modified reflectarray element is characterized by a much more linear phase tunability than the standard element. This is reinforced by data presented in Table 1, which shows mean values of phase-shift errors for the retransmitted signal with reference to an ideal phase shifter. The errors were calculated using Eq. 2:

where:

β = the phase constant of the propagation constant in the transmission line;

li = the geometric length of the transmission line;

φi = the phase shift calculated using the WGS method; and

N = the number of calculation points.

The data in Table 1 indicate that the modified reflectarray element shifts the phase of the retransmitted signal in a manner very close to an ideal phase shifter. The mean phase-shift error, as defined by Eq. 2, is at least two times lower for the modified element than for the standard reflectarray element. With its increased linearity of phase tenability, the modified reflectarray element is less sensitive to manufacturing errors and tolerances in transmission line length.25 Owing to the high linear phase tunability, it is not necessary to analyze the retransmitted signal for every transmission line length in the modified reflectarray antenna. Figures 6(b) and (d) show that the attenuation of the retransmitted signal is approximately 0.05 dB higher for the modified reflectarray element than for the conventional one.

6. These plots show (a) the phase shift and (b) attenuation of the retransmitted signals for the reflectarray element of Fig. 2(a), and (c) the phase shift and (d) attenuation of the retransmitted signals for the modified reflectarray element for a plane wave, illuminating from θ = 23 deg.

To verify the simulation results, a version of the modified aperture-coupled antenna was manufactured using RT/duroid® 5880 printed-circuit-board (PCB) material from Rogers Corp., with a dielectric constant of 2.20 in the z direction (thickness) at 10 GHz and ROHACELL HF foam (Fig. 7). Comparing the measured results (Fig. 8) with the simulations [Fig. 4(a)] reveals that the VSWR of the physical antenna is very close to that of the simulated antenna.

7. The photographs show (a) the assembled test module and (b, c, d) circuit boards used in the conventional and modified antennas.

To verify that the modified antenna design can be used as a reflectarray element, a Waveguide Simulator sample of the element was also manufactured (Fig. 9). It consists of two reflectarray elements in the window of a WR-430 waveguide. The measurement system includes a PNA series microwave vector network analyzer (VNA) from Agilent Technologies, a coaxial-to-waveguide adapter, and waveguide transitions. The WR-430 waveguide dimensions determine the direction of a virtual plane wave impinging upon the reflectarray elements. Using Eq. 1, it is possible to calculate the angle of the plane wave propagation: θ0 = 23 deg.

8. These VSWR measurements of the modified aperture-coupled antenna were made with a commercial VNA.

9. These photographs show (a, b, c) the modified aperture-coupled reflectarray element based on the Waveguide Simulator sample and (d) the test setup used to evaluate the circuits.

To evaluate the physical unit, the lengths of the transmission lines in both reflectarray elements were changed simultaneously, and the reflected signals measured. The measured results in Fig. 10 show the high linearity of the modified element’s phase tenability, which is in agreement with the simulation results [see Fig. 6(c)]. The average retransmitted signal loss amounts to 0.20 dB at 3.4 GHz and is approximately 0.05 dB higher than its simulated value, perhaps due to the higher losses exhibited by an actual circuit substrate versus the model.

10. These plots show measured (a) phase shift and (b) loss of the retransmitted signal based on the Waveguide Simulator method.

The phase-shift errors of the reflectarray units were also compared with an ideal phase shifter, as calculated by Eq. 2, and shown in Fig. 10(b) and Table 2. Again, the measurement results are in very close agreement with the simulation results, with only slight differences. These differences may be due to nonuniform shortening of the transmission lines in reflectarray elements.

In summary, the modified microstrip aperture-coupled antenna and reflectarray element delivers broader bandwidth and higher efficiency than a conventional unit, with high linear phase tunability versus transmission line length. This modified element shows great promise for designing broadband reflectarray antennas with outstanding performance for a variety of commercial and military applications.

Download this article in .PDF format
This file type includes high resolution graphics and schematics.

Michal Zebrowski, Microwave Engineer, M.S.

Bumar Elektronika SA, Poligonowa 30 St., 04-051, Warsaw, Poland; (+48) 22 48 65 475, www.bumar.com/elektronika.

References

1. F. Venneri, et al., “Experimental Evaluation of the Phase of the Field Scattered by Microstrip Patches for Reflectarray Design,” Microwave and Optical Technology Letters, Vol. 34, No. 3, August 2002, pp.163-164.

2. J. Huang and J. Encinar, Reflectarray Antennas, Wiley Interscience, New York, 2007.

3. J. Huang, “Bandwidth Study of Microstrip Reflectarray and a Novel Phased Reflectarray Concept,” Antenna and Propagation  Society International Symposium 1995, AP-S Digest, 1995, pp. 582-585.

4. J. Encinar and A. J. Zornoza, “Broadband Design of Three Layer Printed Reflectarrays,” IEEE Transactions on Antennas and Propagation, Vol. 51, No. 7, July 2003, pp. 1662-1664.

5. David Pozar, “A review of Aperture Coupled Microstrip Antennas: History, Operation, Development and Applications,” ECE, University of Massachusetts, May 1996.

6. Martin Sayeed, “A Design Approach Analysis for inset fed microstrip antenna,” International Conference on Circuits, Systems, Signal and Telecommunications, CISST 2010.

7. Raj Garg, et al., Microstrip Antenna Design Handbook, Artech House, London, 2001, Chap. 9.5.1.

8. T. Milligan, Modern Antenna Design, Wiley Interscience, New York, 2005, Chaps. 6.9 and 6.3.

9. K. Wong, Compact and Broadband Microstrip Antennas, Wiley Interscience, New York, 2002, Chap. 3.4.

10. J. Volakis, Antenna Engineering Handbook, 4th ed., McGraw-Hill, New York, 2009, Chap. 16.9.

11. V. Srinivasan, et al., “Multiport Network Model for Chip Resistor Loaded Rectangular Microstrip Antennas,” Microwave and Optical Technology Letters, Vol. 24, No. 1, 2000, pp. 11-13.

12. G. Kumar and K.P. Ray, Broadband Microstrip Antennas, Artech House, Norwood, MA, 2003, Chaps. 3, 4.3, 6.5, and 6.6.

13. K. Wong and W. Hsu, “A Broadband Rectangular Patch Antenna with a Pair of Wide Slits,” IEEE Transactions on Antennas and Propagation, Vol. 49, No. 9, September 2001, pp. 1345- 1347.

14. J. Zuercher and Fred Gardiol, Broadband Patch Antennas, Artech House, Norwood, MA, 1995, Chaps. 2.5.3 and 3.

15. Evonik Industries AG, 2012, www.rohacell.com.

16. J.R. James and P.S. Hall, Handbook of Microstrip Antennas, Peter Peregrinus, London, 1989, Chap. 1.2.

17. S.D. Targonski, et al., “Wideband aperture coupled microstrip patch array with backlobe reduction,” IEE Electronics Letters, Vol. 33, No. 24, November 1997, pp. 2005-2006.

18. P.W. Hannan and M.A. Balfour, “Simulation of a Phased-Array Antenna in Waveguide,” IEEE Transactions on  Antennas and Propagation, Vol. 13, No. 3, May 1965, pp. 342-353.

19. R.C. Hansen, Phased Array Antennas, 2nd ed., Wiley, New York, 2009, Chap.15.3.

20. H. Visser, Array and Phased Array Antenna Basics, Wiley, New York, 2005, Chap. 11.4.

21. Merrill Skolnik, Radar Handbook, 2nd ed., McGraw-Hill, New York, 1990, Chap. 7.4.

22. S.D. Targonski, et al., “Design of Wideband Aperture Stacked Patch Microstrip Antennas,” IEEE Transactions on Antennas and Propagation, Vo. 46, No. 9, September 1998.

23. David M. Pozar and S.D. Targonski, “Improved Coupling for Aperture Coupled Microstrip Antennas,” Electronics Letters, Vol. 27, No. 3, June 1991.

24. H. Rajagopalan and Y. Rahmat-Samii, “On the Reflection Characteristics of a Reflectarray Element with Low-Loss and High-Loss Substrates,” IEEE Antennas and Propagation magazine.

25. N. Misran, “Design optimization of ring elements for broadband reflectarray antennas,” IEEE Proceedings on Microwave, Antennas  and Propagation, Vol. 150, No. 6, pp. 440-444.

Download this article in .PDF format
This file type includes high resolution graphics and schematics.