Designers of third-generation (3G) wireless handsets can take several different approaches to meeting multimode, multiband front-end radio requirements.
Third-generation (3G) multimode wireless communications promise mobile multiedia services as a result of enhanced data rates. For example, High Speed Packet Access (HSPA) technology is providing evolutionary performance improvements through Universal Mobile Telecommunications Systems (UMTS) networks. High Speed Downlink Packet Access (HSDPA) is delivering faster downlink speeds and user data throughput of 1 Mb/s while High Speed Uplink Packet Access (HSUPA) is paving the way for mobile interactive applications and services. Of course, for handset suppliers to keep pace, they count on highly integrated front-end integrated circuits (ICs) that are capable of supporting multiple frequency bands with high quality and increased battery life. For handset designers, numerous solutions exist; two possible approaches include transmit modules with integrated power amplifier (PA), filters, and duplexers and a more discrete solution which includes broadband power amplifiers and a power-management IC (PMIC).
In a typical 3G multimode front-end architecture (Fig. 1), key RF function blocks include transceivers, transmit surface-acoustic-wave (SAW) filters, power-amplifier (PA) modules for GSM/EDGE and UMTS, duplexers, isolators, and a front-end frequency-conversion module. Some handset designs do not include isolators but can pay a price in performance under different load mismatch conditions. To enhance transmitter efficiency, some handsets include a PA power-management IC. It is an additional component that adds cost, but provides benefits that far outweigh its added size and cost.
For GPRS/EDGE, the industry is witnessing a migration from PA modules to transmit modules that include harmonic filtering and antenna switches. The reason for the migration is the resulting reduced component placement, development time, and solution yield through higher levels of integration. In UMTS, however, even though there is a trend to higher levels of integration with PAs, filters, and duplexers, a strong need remains for platform flexibility in order to optimize performance. These demands require an alternate approach to address key concerns for handset designers.
Worldwide UMTS network deployment today shows uplink frequency usage dominated by several frequency ranges: Band I uplink, from 1920 to 1980 MHz; Band II uplink, from 1850 to 1910 MHz; and Band V uplink, from 824 to 849 MHz. There is also growing demand for operation in UMTS Bands IV, VI, VIII, IX, and X, with plans to deploy Bands III and VII at a later stage. Producing a platform to address all these bands would require several PAs or transmit modules optimized for each band. This impacts development time by having to replace key components, such as the PA, for each regional application variant. Re-optimization of impedance matching, and control interfaces creates learning curves with each new product. In addition, there is the lack of flexibility with the highly integrated approach incorporating filters and duplexers.
The transmit-module approach offers distinct advantages in terms of reduced component placements and size for single- band applications. Considering PA and duplexer interface issues, there is also the benefit of reduced design cycle time. Transmit module solutions from RFMD include models RF6241, RF6242, and RF6245 for UMTS bands of I (1920 to 1980 MHz), II (1850 to 1910 MHz), and V (824 to 849 MHz), respectively. These transmit modules include transmit SAW filters and duplexers and have the additional benefits of reduced current consumption for backed-off (linear) power levels using a digitally selected lowpower mode (Fig. 2).
As the industry migrates to dualband and tri-band multimode 3G handset solutions, almost mirroring the evolution of 2G, region-specific transmit module with its clear benefits for singleband applications yield to the flexibility of more discrete solutions. An alternate approach to implementing a multiband front end is the use of a broadband PA solution, which can be used across several bands. Model RD6280 is a frontend module consisting of three components: a model RF6285 broadband PA, RF6281 Band I and II optimized singlepath PA, and model RF6280 powermanagement IC. It addresses not only various dual- and tri-band configurations, but single-band applications where the priority is for performance rather than size and/or cost (Fig. 3). Such a discrete solution retains flexibility in the filter supply chain.
Current consumption is another concern for UMTS handset designers. Additional features attractive to handset consumers also compete for talk time. Handset owners want smaller and lighter phones, which limits the size and capacity of the battery pack. Since the PA can account for as much as 40 percent of power consumption, a major focus is being placed on efficiency improvements in the PA solution. Although a PAs peak efficiency is important, it is only one factor in determining talk time for a handset design. In a comparison of two power usage profiles for typical cellular networks (Fig. 4), statistically a handset operates primarily in the power range between -15 and +10 dBm. Based on network operator feedback, there is growing consensus among the design community that performance at backed-off power levels is more important than at peak power.
There are several approaches to improve the PA performance at backedoff power levels. The technique employed in the RF6241 transmit module is a combination of power mode selection and analog bias control. At +18 dBm, the amplifier is switched to a low-power mode, with further current reductions coming from continuous analog bias adjustments across the lower power levels. A 70-percent reduction in low-power current consumption can be achieved using the analog bias control (Fig. 5). In the model RD6280 IC, a power-management DCto- DC converter is adjusted at each power level in conjunction with the analog bias control, providing more than 80- percent current reduction at low-power levels. The RF6280 was specifically designed for optimized efficiency based on the light load condition of the RF6285 under low-power operation. Analog bias control in combination with collector control is a technique proven in mass production and employed by several Tier 1 handset manufacturers.
The RF6280 DC-DC converter section operates as a pulse-width-modulated (PWM) voltage mode controller which has a transfer function of 2.5 times the Vset voltage out = 2.5(Vset)>. By using the DC-to-DC converter to optimize the PA collector voltage for each power level, the battery current consumption can be reduced by almost the ratio of Vout to Vin. Using the equation for the DCto- DC converter, Pout = ?Pin, where ? is efficiency, the equation for battery current (Ibat) can be derived as Ibat = (Vout /?Vin) Iload where Iload is the current of the load. This equation shows that the battery current is the ratio of Vout to Vin, assuming 100-percent converter efficiency and a constant load across all operating voltage levels.
Fig. 6 illustrates how the use of power management reduces battery current consumption. It shows measured battery current consumption when the PA collector voltage is optimally adjusted for each power level. The data shows that at 0 dBm, the battery current consumption can be reduced by about 79 percent compared to that of the PA alone with no collector voltage adjustment. Another way to reduce battery current consumption is by employing analog bias control techniques. The RF6285 and RF6281 PAs are designed with a bias circuit that allows for the control voltage (Vctrl) to be adjusted so that the PA can be biased lower at lower power levels. By adjusting Vctrl, the battery current consumption can be reduced by about 48 percent compared to that of a PA with no base-bias change. The "base-bias adjust" curve in Fig. 6 shows the result of changing the PA base bias only. In a way to further reduce battery current consumption, both the PA base bias and collector voltage in combination can be reduced at lower power levels. The resulting effect on battery current as shown by the "base-bias adjust + DC-DC" curve in Fig. 6 is about an 88-percent reduction in current. Using this combination of control voltages, the total battery current drawn by the PMIC and PA will be less than 7 mA at amplifier output-power levels below 0 dBm.
The peak-to-average power ratio of a voice-modulated signal is less than that of an HSDPA-modulated signal. Since the handset must operate in both modes, the PA must be biased high enough to maintain sufficient linearity to achieve system adjacent-channelleakage- ratio (ACLR) requirements. Once the PA bias voltages are set for HSDPA requirements, these same voltage levels provide additional ACLR margin under voice operation. Since this additional ACLR margin is not necessary, ACLR performance can be traded off for additional battery current reduction. Fig. 7 illustrates that at the highest power level, 30 mA current savings can be achieved in voice operation by reducing the control voltage settings from that used in HSDPA operation. This flexibility gives the designer the handles to trade-off linearity margin, efficiency, and power depending on the mode of operation.
Thermal dissipation in the handset continues to be a concern for designers. A significant portion of this dissipation is due to the power amplifier. The use of a DC-to-DC converter will allow the PA collector voltage to be operated at the minimum specified voltage at maximum output power independent of the battery voltage. To illustrate reductions in dissipated power achieved using DC-to- DC converters, a simple example is shown using the RF6285 operating at maximum output power.
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The RF6285 will typically draw 450 mA of current when operating at +27- dBm output power. From the equation showing dissipated power as a function of input power and output power, Pdiss=Pin -Pout, Pin will determine how much power will be dissipated in the PA since Pout will be constant at +27 dBm (0.5 W). For the case of using the PA with a fully charged battery with no power management, the Pin value would be: Pin = 4.5 V X 0.45 A = 2.03 W. In this case, the dissipated power would be 1.53 W. In the case where power management is used, the PA collector will be set to 3.1 V, so Pin = 3.1 V X 0.45 A = 1.39 W and the power dissipation would now be only 0.89 W. This drop in voltage results in about a 42-percent reduction in PA power dissipation which has been shown to reduce the PA die temperature by as much as 42°C.
Handset field performance is of great importance to operators and handset manufacturers. EVM and ACLR system requirements, as well as total-radiated- power (TRP) and specific-absorption- rate (SAR) limits necessitate robust performance under mismatched load conditions. Additionally, post calibration handset efficiency and power accuracy can be negatively impacted. PA load sensitivity contributes significantly t the overall performance of the transmit chain. In order to reduce PA load sensitivity, a quadrature architecture is used in the RF6281, RF6285, RF6241, RF6242, and RF6245 PAs. They are designed using parallel path amplifiers with lead/lag splitter and combiner networks. In use, a wireless network may create amplifier paths that are 90 deg. out of phase relative to each other, requiring robust operation from the PAs.
Fig. 8 illustrates what happens when a mismatch is applied to the output of the lead/lag combiner network. In this simulation, a 5.0:1 VSWR mismatch is applied to the Port 1 combiner output. A 5.0:1 VSWR was used because that is the expected worst-case mismatch that would occur when placed in a handset environment. The effect at Ports 2 and 4 is an impedance transformation which presents a 2.0:1 VSWR to each PA output. Front-end component insertion losses provide some isolation from antenna mismatch. Typical front-end component insertion losses are at least 3 dB which would limit the worst case VSWR at the lead lag combiner output at about 3.0:1, rather than 5.0:1.
Fig. 9 shows the RF6285´s measured ACLR performance versus VSWR to demonstrate the benefits of the PA´s quadrature architecture. As a reference, at the worst-case VSWR of 3.0:1, the ACLR degradation is less than 2 dB compared to 10 dB for a single-ended PA. For the worst-case 3.0:1 VSWR, the power degradation of the RF6285 is only 2.5 dB, compared to a degradation of 3.5 dB for a single-ended PA at the same VSWR (Fig. 10). Fig. 11 shows the peak collector current performance of the RF6285 versus VSWR. At the worst-case VSWR of 2.0:1, peak collector current for the RF6285 increased by only 3 percent compared to a singleended PA, which has suffered increases in collector current by as much as 20 percent under the same worst-case VSWR conditions.
Designers of 3G handsets face numerous challenges, with priorities set by final applications. Solutions like the RD6280 module provide the benefit of flexibility for multiband applications while maintaining optimum performance across power levels and operating modes. In contrast, highly integrated solutions like the RF6241, RF6242, and RF6245 devices address the need for reducing component count. Both solutions address the need for improved current consumption at backed-off power levels, with RD6280 offering more performance optimization with the inclusion of a PA PMIC. The RD6280 and RF6241 devices are sampling to alpha customers now, with the RF6242 and RF6245 sampling first quarter of 2008. RF Micro Devices, Inc., 7628 Thorndike Rd., Greensboro, NC 27409-94219; (336) 664-1233, FAX: (336) 931-7454, Internet: www.rfmd.com.
