1. These drawings show the configuration of the four-way power divider: (a) the middle input layer, (b) the second and fourth ground layers, (c) the top and bottom output layers, and (d) the view of the overall power divider.
Zheng Zhang, Yong-Cheng Jiao, Xiao-Ming Wang, Li-Na Chen, and Fu-Shun Zhang
Power dividers are widely used in RF and microwave circuits and systems to distribute signals for various purposes, such as amplification and transmission. They are employed in amplifiers, mixers, phase shifters, and the feeding network for an antenna array.1,2 In recent years, there has been a great interest in designing microwave circuits with multilayer microstrip technology which can be used to miniaturize power dividersfor example, low-temperature-cofired-ceramic (LTCC) and laminated multichip-module (LMCM) approaches.3,4 In these newer technologies, a common-ground structure is frequently used. The usual method for achieving efficient power transfer between two sides of the common ground plane is to use a wired via; however, this is difficult and labor-intensive to fabricate. Work performed in ref. 5 reported on a broadband wireless via using microstrip-slotline transitions, which were implemented in the studies of refs. 6 and 7.
To demonstrate the effectiveness of miniaturizing RF/microwave power dividers by means of microstrip-slotline transitions, a compact four-way power divider was designed for broadband use from 3.6 to 8.4 GHz. The design was found to exhibit insertion loss of 6.5 dB, with insertion loss maintained within a 0.5-dB window. The power divider also achieved high isolation between the output ports and better than10-dB return loss at the input and output ports.
2. These plots show the simulated S-parameters for the power divider with and without air gaps, which impact impedance matching.
Figure 1 shows the four-substrate construction of the multilayer four-way power divider. The circuit design consists of five conductor layers interleaved by four dielectric layers. The four dielectric substrates are the same size of 14 x 16 mm, with thickness (h) of 0.6 mm relative dielectric constant, εr, of 2.65, and dielectric loss tangent (tan δ) of 0.002. The input port is at the middle layer, while the four output ports are formed at the top and bottom layers of the multilayer structure. The ground planes, which have coupling slots in the shape of a narrow rectangle ending with two circles, are at the second and fourth layers of the structure. Due to the symmetry of the multilayer power divider, the RF signals at the input port are equally divided in amplitude and phase among the four output signal ports.
Figure 1(a) shows the middle metalized input patch for the proposed power divider. The dimensions of this T-shaped microstrip structure are determined by the parameters W1, L1, W2, L2, W3, L3, W4, and L4 in Fig. 1(a). These two impedance transforms, with dimensions of W2 x L2 and W3 x L3, were introduced to improve the impedance bandwidth of the power divider.
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Figure 1(b) shows the configurations of the second and fourth layer grounds, which are the same. The length of the slot line, W5, is chosen to be one-quarter of the effective wavelength, λeff, at the center frequency (6 GHz). The effective wavelength, λeff, can be expressed as:
where:
c = the velocity of light in a vacuum,
f = the center frequency of the power divider, and
εeff = the effective dielectric constant of the transmission medium.
For a typical slotline, both of the regions above and below the substrate are air, and the effective dielectric constant of the transmission medium is the average value of the air and the substrate.8 That is:
In this case, for the slotline used in the proposed power divider, the regions above and below the substrate are not air but other substrates, with the same permittivity. As a result, the effective relative permittivity used in Eq. 1 should be calculated by Eq. 3:
To achieve a higher return loss at the microstrip ports, the slot width L5 should be chosen for an impedance close to 50 Ω as seen from the microstrip side. As detailed in refs. 9 and 10, the impedance of the slot as seen from the microstrip line, Zin, should equal n2Zs, where the impedance transformation ratio n and the slot impedance Zs are determined by the slot width and substrate characteristics. In the present design, the slot width is chosen to be Zs = 95 Ω and n = 0.8, so that the impedance of the slot as seen from the microstrip line Zin = (0.8)2 x 95 = 60.8 O. The radius of the slot circle, r1 , is selected to be between one and two times of L5, since it is used to terminate a slot line.
Figure 1(c) depicts the same four output metalized patches. The radius r2 of the quarter-circle is also selected to be a quarter-wavelength at the center frequency (6 GHz). The widths of microstrip line output and input ports are both W3. The final values of all variables in Fig. 1 are W1 = 8 mm, W2 = 0.4 mm, W3 = 2 mm, W4 = 1.64 mm, W5 = 7.86 mm, W6 = 5 mm, W7 = 6 mm, L1 = 3.75 mm, L2 = 0.2 mm, L3 = 1.5 mm, L4 = 6.5 mm, L5 = 0.35 mm, L6 = 0.8 mm, L7 = 8.44 mm, r1 = 1 mm, and r2 = 6 mm.
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3. These plots show the simulated and measured S-parameters for the four-way power divider.
Because the proposed power divider was implemented using four separate dielectric substrates, substrate alignment was a concern. For example, air gaps between different layers might be introduced during the fabrication and impact the impedance matches at those areas. These air gaps may cause undesirable influence on the S-parameters. Suppose the four substrates are interleaved by three 0.1-mm-thick air gaps. Figure 2 shows the simulated S-parameters of the proposed power divider with and without the three air gaps. It was found that the introduced air gaps have little effect on the return loss of port 1, but they significantly affect the insertion loss. A mismatch between the input and output ports induced by the introduced air gaps, which lower the effective relative permittivity of the slotline, increases the insertion loss significantly. Thus, it will be much better to implement this power divider in LTCC technology for its advantage of eliminating much of the substrate alignment tolerance.
To understand the behavior of the proposed power divider and obtain the optimum performance parameters, a simulation was performed with Microwave Studio from CST, and measurements were made with a model 37269A vector network analyzer (VNA) from Anritsu Company. Because of the limited technology level available for fabrication, the proposed power divider was designed by using four separate dielectric substrates rather than LTCC technology.
Figure 3(a) presents the simulated and measured return and insertion losses. From the results, it can be seen that the simulated and measured results are in close agreement. The discrepancies result most likely from substrate alignment tolerances and the electrical effects of the SMA connectors added to the power divider, which were not incorporated into the simulations. Over the full 3.6-to-8.4-GHz operating band, the return loss is better than 10 dB, and the insertion loss is equal to 6.50.5 dB, indicating that the designed power divider can successfully divide an incoming signal into four signal paths with equal amplitude at each output. The simulated and measured isolation results are plotted in Fig. 3(b). It can be seen that the isolation is better than 10 dB.
The four-way power divider demonstrated here shows reasonable performance even when fabricated with four separate laminate boards, rather than a more appropriate technology such as LTCC. The approach is suitable for miniaturizing a power divider while maintaining low insertion loss, good amplitude balance, and high isolation between ports. The approach is well suited for a variety of C-band applications requiring low loss and small size.
Acknowledgment
This work was supported by the RIM Advanced Antenna Research Program.
References
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