Bandpass Filter Passes 2.4- and 5.2-GHz Bands

In another approach,³ two homocentric embedded defected ground waveguide (DGW) resonators were adopted for a dual-band filter design, each fed by a 50-Ω microstrip line. In another work,^4,5 a dual-band bandpass filter with meander-loop resonator and complementary split-ring resonator defected ground structure (SRR DGS) was proposed. Finally, an aperture compensation technique was proposed for enhancing the coupling in the parallel-coupled microstrip lines employed in these filters,⁶ although there is yet little research available on combining dual-band filter designs with the aperture compensation technique.

For improved performance in dual-band communications applications, a dual-band bandpass filter (BPF) was developed. It features a dual-mode resonator and SRR DGS excited with common improved input/output (I/O) coupling structure using an aperture compensation technique. The improved I/O structure is designed to provide greater coupling than traditional coupling structures without adding size. The performance of the improved coupling structure was confirmed by means of two-port equivalent-circuit analysis. The coupling structure and filter achieves high selectivity by the use of several finite attenuation poles within the stopbands.

This new design approach also offers a great deal of flexibility in the manner of independently controllable center frequencies within the passbands and bandwidths that can be tuned over a wide range. The filter structure is relatively simple and compact. The design approach was applied to the development of a dual-band filter for use at the wireless-local-area-network (WLAN) frequency bands of 2.4 and 5.2 GHz.  

The improved dual-mode resonator structure is shown in Fig. 4(a). The dual-mode characteristics are realized by loading a stepped-impedance-line stub at the center of the open-loop resonator. Since the resonator is symmetrical to the T-T' plane, the odd- and even-mode method can be applied to analyze it. The equivalent circuit of Fig. 4(b) can be applied for analysis of the odd-mode excitation. The one-port input impedance of that equivalent circuit can be found with the aid of Eq. 2:

Z_in,odd = jZ₀tanθ₀   (2)

From the resonant condition that Y_in,odd = 0, the first odd-mode resonant frequency can be determined by means of Eq. 3:

f_odd = c/[4(2a – g/2)(ε_eff)^0.5)]   (3)

where:

c = the speed of light in free space, and

ε_eff = the effective dielectric constant of the substrate material.

For an even-mode excitation, the equivalent circuit of Fig. 4(c) should be applied. Its one-port input impedance can be found by means of Eqs. 4-6: 

Z_in1 = -j(1/ Z₂tanθ₂)   (4)

Z_in2 = Z₁[(Z_in1 + jZ₁tanθ₁)/(Z₁ + jZ_in1tanθ₁)]   (5)

Z_{in, even} = Z₀[(Z_in2 + jZ₀tanθ₀)/( Z₀ + jZ_in2tanθ₀)]   (6)

From the resonance condition that Z_in,even = 0, it is possible to achieve the equality of Eq. 7:

R₁R₂tanθ₀tanθ₁tanθ₂ - R₁tanθ₀ – tanθ₁ - R₂tanθ₂ = 0   (7)

where:

R₁ = Z₁/Z₀, and R₂ = Z₁Z₂.

Under the special condition that Z₀ = Z₁ = Z₂, the first resonant frequency can be found by applying Eq. 8:

f_even = c/[2(2a – g/2 + b + c)(ε_eff)^0.5]   (8)

Equations 3 and 8 show that changing the values of b and c only affects the even-mode resonant frequency, leaving the odd-mode resonant frequency unchanged. Knowing this, by adjusting the values of b and c, the even-mode resonant frequency can be tuned while also adjusting the center frequencies and bandwidths.

Figure 9 is a photograph of the fabricated dual-band filter. Simulations and measurements were accomplished by means of the High Frequency Structure Simulation (HFSS 13.0) simulation software and model E5071C microwave vector network analyzer (VNA), respectively, both from Agilent Technologies (now Keysight Technologies).

Figure 10 presents simulated and measured results for the proposed filter. The two filter passbands are centered at 2.53 and 5.11 GHz, with minimum insertion losses of 1.1 and 3.0 dB, respectively, for the two passbands. The return losses within the two passbands were better than 14 dB. Two attenuation poles were realized for the dual-band BPF structure, at 2.83 and 6.08 GHz, which helped to greatly improve the selectivity of the proposed dual-band BPF. As can be seen from the plots, the measured results mainly agree with the simulated performance values. Discrepancies between the measured and simulated performance levels (Fig. 8) can be attributed to fabrication tolerances in realizing the dual-band filter on commercial microwave circuit substrate material.

Wu Guo-An, Professor

Zhang Bi-Xia, Master’s Degree Candidate

Xu Qin-Fen, Lecturer

Li Wen-Guang, Lecturer

Institute of Microwave Technology Application, Huazhong University of Science and Technology, Wuhan, 430074, People’s Republic of China